DC-Controlled Encoding For Optical Storage System

ABSTRACT

A loss of performance of slicer adaptation at high capacities due to the mismatch between the exact bits used in the computation of the RDS for the DC-control on the one hand and the often erroneous threshold decisions that are preliminarily made based on the HF waveform on the other hand, is resolved by performing a new method of DC-control at the encoder: the RDS is modified such that it is not based on the exact channel bits, but on the threshold decisions from a synthetic HF signal waveform that is generated based on a nominal MTF (modulation transfer function) or its IRF (impulse response function) of the channel. In this way, the impact of the erroneous threshold decisions in the receiver are already taken into account at the encoder, and the slicer control is no longer negatively affected thereby.

This invention relates to a method and apparatus for creating asubstantially DC-controllable code representative of a modulated signalreceived via a channel, in which a running digital sum is computed usinga plurality of bits or bit-estimates. The invention also relates to anencoder incorporating such a method or apparatus, and an optical datastorage system including such an encoder and a receiver.

Optical data storage systems provide a means for storing greatquantities of data on a disc. As is well known in the art, an opticaldisc comprises at least one track which is capable of containing datawritten therein. The disc may be embodied so as to be a read-only disc:the disc is manufactured with data recorded in the track, and this datacan only be read from the disc. However, recordable and (re-) writeableoptical discs, allowing a user to record data on a disc, are also known;in this case, a disc will normally be manufactured as a blank disc, i.e.a disc having a track structure but without data recorded in the track.Similarly, disc drives may be designed as read-only devices, i.e.devices only capable of reading information from a recorded disc.However, disc drives may also be designed for writing information intothe track of a recordable disc.

Physically, the information bearing portion of an optical disc is aseries of pits, or bumps, arranged to form a spiral track. Data isencoded in the length of the individual pit-marks (or pits) and of thespaces (also known as land-marks) between pits. A laser beam reflectedfrom the optical disc is modulated by the pits and spaces, and receivedby a detector which produces a similarly modulated electrical signal, ortrack data signal.

The track data signal is demodulated to recover digital informationstored on the disc by observing the amplitude of the track data signalresponsive to a data clock. The characteristics of the track datasignals enable the data clock to be derived from the track data signalusing phase locked loop (PLL) circuitry. Data is encoded such that ifthe amplitude of the track data signal is approximately the same fromone sample to the next, the corresponding bit has a value of ‘0’, and avalue of ‘1’ otherwise.

Referring to FIG. 1 of the drawings, a receiver with advanced signalprocessing and bit-detection for an optical storage system isillustrated schematically. The light of the laser beam that is reflectedand diffracted by the pit-structures on the disc, is captured by aphoto-detector, which generates an electrical signal, often called RF orHF signal. An A/D converter generates digital samples of the(electrical) signal waveform. Those samples are corrected for aDC-offset by a slicer. The origin of the DC-offset can be due tofingerprints and other (low-frequent) disturbances on the disc. Thesliced HF samples enter a first sample-rate converter, that has clockinformation derived from a wobble signal (as is available for arecordable/rewritable disc). Subsequently, the HF-signal samples areequalized (by a linear equalizer, and possibly followed by a non-linearequalizer known as Limit-Equalizer), after which the samples enter ajitter-based PLL with a second sample-rate converter. These outputsamples are bit-synchronous and are used for a runlength-pushbackbit-detector (which is used as a preliminary bit-detector), the outputof which is coupled to an adaptive look-up table of reference levels(RLU), that are used in the Viterbi bit-detector (VD). Moreover, thebit-synchronous HF-samples are further used to drive the slicer-modulethat corrects for the DC-offset in the signal.

Thus, during readout, the laser beam from each spot is diffracted by the2D pattern on the disc, which is received as high-frequency (HF) signalwaveforms. HF represents the intensity of light detected within thecentral aperture in the exit pupil plane of the objective lens. As thedensity of recorded information increases, the physical proximity of onerecorded bit to the next adjacent bit tends to cause interferencebetween adjacent bits, and the HF value is influenced by thisInter-Symbol-Interference (ISI). The ISI increases with increasingbit-density along the track-direction.

It is well known to encode the digital data so that only specifiedsequences of transitions of the medium are permitted. These permittedsequences are known to reduce the effects of inter-symbol interference.Furthermore, these constraint sequences are characterized by a minimummark-size (or pit-size) that is larger than in the unencoded case;therefore, for a write-channel that is limited by the smallest mark thatcan be written, the use of such a code generating these constaintsequences is also advantageous. In particular, it is common to userun-length-limited (RLL) encoding of the digital data to generate anRLL-encoded stream of bits to be read by read channel digital processingmeans. This encoded stream of bits is often referred to as channel bitsin that they represent the stream of bits encountered by read channelcomponents of the device. In any event, an RLL (d,k) code is generallyused to encode an arbitrary set of data bits into a stream of channelbits such that the encoded channel bits satisfy the “d” and “k”constraints, and a wide variety of encodings that satisfy a given (d,k)constraint may be used. Several such encodings are well known in thefield of optical data storage.

The linear density of the bits and the track pitch are fixed by thespecification of the particular optical disc format. For example, CDdiscs employ a track pitch of 1.6 μm having approximately 80 thousandchannel bits per linear inch, while DVD employs a track pitch only aboutone-half as wide (0.74, μm) and having approximately 200 thousandchannel bits per linear inch. In high capacity optical storage systems,beyond the Blu-ray Disc (BD) capacity of 23-25-27 GB, many aspects ofthe (already) advanced receiver illustrated schematically in, anddescribed with reference to, FIG. 1 cannot be used in their existingform. In particular, at capacities beyond 31 GB, there are verysignificant changes in the characteristics of the signal waveform. Forexample, assuming a track pitch always equal to 320 nm, the tangentialbit length for 31 GB becomes T=60.5 nm, so that the highest frequency inthe system (with d=1 coding), which is equal to ¼T (which equals 4.13:m⁻¹), becomes very close to the cut-off frequency of the channel, givenby:Ω_(c)=2NA/λ  (1)which equals 4.20: m⁻¹ for Blu-ray Disc (BD) parameters. The gradualdegradation of the HF signal waveform and the eye pattern going from BD23 GB, over BD 27 GB and BD 31 GB, up to BD 35 GB can be seen in FIGS. 2and 3 of the drawings.

At these high densities beyond BD 31 GB, the 2T-runs are not well, oreven not at all, transferred by the optical channel. Typical effects canbe seen in FIGS. 4 and 5 of the drawings. FIG. 4 illustrates (around bit40) 2T runs (of pit-type) alternating with 3T runs (of land-type), withas a result, a waveform of low amplitude, but with the signal alwaysabove the slicer level, so that the 2T runs are “lost” for a simplethreshold detector. FIG. 5 illustrates 2T runs in the vicinity of longruns of opposite polarity (opposite to that of the 2T run): in the HFsignal, these 2T runs may disappear to a large extent, and result inonly a small indentation of the waveform that could be interpreted by asimple threshold detector as a much longer run (of the same polarity).

Although for bit-detection for the BD-system, use is made of the moreadvanced Viterbi detection (also known as PRML detection, standing forpartial-response maximum-likelihood), for the control of the DC-offset(or slicer-level) in the “slicer” module of FIG. 1, a simple thresholddetection is still being used.

Thus, at higher capacities (say, for BD>27 GB), where the highestfrequencies in the system are around or beyond the cut-off frequency ofthe channel's modulation transfer function (MTF), which characterisesthe model of an optical channel, leading to very poor or even notransfer of the high frequency information, simple threshold decisionson the HF signal waveform result in many decision errors, which increasethe standard deviation of the slicer level updates, which necessitatesthe reduction of the bandwidth of the slicer-control, and consequentlyprohibits the slicer control for being as fast as it is for lessaggressive capacities (where threshold bit-decisions are of much betterquality). It is an object of the present invention, therefore, toprovide an encoder for a receiver for an optical data storage system,and a receiver and optical data storage system including such anencoder, arranged to overcome the above-mentioned problems that aretypical of a high-capacity optical storage system, and enable the slicercontrol to be performed at a faster rate relative to prior artarrangements that use simple threshold bit-decisions on the signalwaveform.

Thus, in accordance with the present invention, there is providedapparatus for creating a substantially DC-controllable channel bitstreambased upon a channel code, representative of a modulated signal receivedvia a channel, the apparatus comprising means for receiving datarepresentative of a nominal modulation transfer function or impulseresponse function of said channel, means for generating a synthetic highfrequency signal waveform using said nominal modulation transferfunction or impulse response function, means for performing thresholddetection in respect of said synthetic high frequency signal waveform toproduce intermediate channel bits, and means for computing a runningdigital sum using said intermediate channel bits.

Also in accordance with the present invention, there is provided amethod of creating a substantially DC-controllable channel bitstreambased upon a channel code, representative of a modulated signal receivedvia a channel, the method comprising receiving data representative of anominal modulation transfer function or impulse response function ofsaid channel, generating a synthetic high frequency signal waveformusing said nominal modulation transfer function ir impulse responsefunction, performing threshold detection in respect of said synthetichigh frequency signal waveform to produce intermediate channel bits, andcomputing a running digital sum using said intermediate channel bits.

It will be appreciated that the running digital sum on said intermediatechannel bits can be used to steer decisions for DC-control to be takenat certain DC-control points in the bitstream (either in the userbitstream or the channel bitstream, depending on the method used tocontrol the running digital sum).

The present invention further extends to an encoder including theapparatus defined above, and to an optical data storage system includingsuch an encoder and a receiver having slicer apparatus for performingthreshold detection in respect of a high frequency signal waveformderived from a modulated signal to create a digital signalrepresentative thereof. It will be appreciated that the digital signalrepresents the as-detected intermediate channel bitstream, which canthen be used for the adaption of the DC-offset or slicer-level in theslicer-module.

In a preferred embodiment of the invention, the code representative ofthe modulated signal is a run length limited (RLL) code and, as such,the apparatus preferably includes means for performing RLL encoding inrespect of a plurality of user bits received via said channel,preferably prior to computation of the running digital sum.

Means are beneficially provided for updating a threshold value inrespect of the slicer apparatus in response to changes in said highfrequency signal waveform derived from said modulated signal andthreshold decisions based on said received moduled signal.

These and other aspects of the present invention will be apparent from,and elucidated with reference to, the embodiment described herein.

An embodiment of the present invention will now be described by way ofexample only, and with reference to the accompanying drawings, in which:

FIG. 1 is a schematic block diagram of a “classical” receiver (alreadywith advanced signal processing) for a BD 23-25-27 GB optical storagesystem;

FIG. 2 is a graphical representation illustrating changes in HF signalwaveform and eye-pattern going from BD 23 GB to BD 27 GB;

FIG. 3 is a graphical representation illustrating changes in HF signalwaveform and eye-pattern going from BD 31 GB to BD 35 GB;

FIG. 4 is a graphical representation illustrating typically problematicHF signal waveforms for 35 GB BD, e.g. 2T|2T runs starting at bitposition 15, and a 3T|2T|3T|2T|3T pattern starting at bit position 37;

FIG. 5 is a graphical representation illustrating typically problematicHF signal waveforms for 35 GB BD, e.g. 2T runs between longer runs (atbit positions 1891 and 1899;

FIG. 6 is a schematic flow diagram illustrating DC-control encoding fora segment of a user bitstream according to a prior art approach;

FIG. 7 is a graphical representation of the principle of sliceradaptation based on threshold decisions from HF signal waveform (notethat the step size is significantly exaggerated);

FIG. 8 is a schematic flow diagram illustrating DC-control encoding fora segment of a user bitstream according to an exemplary embodiment ofthe present invention (Note: “Int. Ch. Bits” stands for Intermediatechannel bits);

FIG. 9 is a graphical illustration of the standard deviation of theslicer-level scaled to standard deviation of signal σ_(SL)/σ_(signal)for BD 35 GB and εε=0.001;

FIG. 10 is a graphical illustration of the standard deviation of theslicer-level scaled to standard deviation of signal σ_(SL)/σ_(signal,)for BD 35 GB and ε=0.0001;

FIG. 11 is a graphical illustration of the standard deviation of theslicer-level scaled to standard deviation of signal σ_(SL)/σ_(signal,)for BD 35 GB and ε=0.00001; and

FIG. 12 is a graphical representation illustrating RDS-traces (runningdigital sum) superimposed on signal waveform: on exact bits (curve A)and on threshold bits (curve B)—note that the two RDS-traces divergefrom each other.

Due to the analogue nature of the waveform sensed by a receiver in anoptical data storage system, and due to the inter-symbol interferenceproblems noted above, it has previously been a problem to sense anddecode the encoded user data bits and, as explained above, the problemis particularly exacerbated as storage density increases. In order topartially resolve such problems, it is known to use sequence detectorsto sense particularly expected sequences of pulses rather thanattempting to detect each discrete individual pulse in the sampledwaveform (without using the waveform samples at the neighbouring bits).In particular, a Viterbi sequence detector (VD) is commonly used to readchannels used in optical storage media, so as to sense most likelysequences of encoded channel bits.

The control loop of the slicer level in a receiver (see FIG. 1 of thedrawings) is required to follow relatively slow variations in thecentral aperture HF signal due to fingerprints, drop-outs, etc. Slicercontrol can be performed faster if, instead of making use ofbit-decisions resulting from a Viterbi bit detector, (preliminary)threshold decisions are performed on a sample-by-sample basis. Slicercontrol, in this case of using threshold decisions, requires that thechannel bitstream is DC-free, i.e. that the running-digital-sum (RDS)has a limited variance, and this DC-free property is generally realisedat the side of the Modulation-Code Encoder, traditionally on the channelbitstream, as will be familiar to a person skilled in the art. Thus,slicer adaptation is performed, in accordance with the prior art, usingthe DC-controlled property of the channel bitstream. A schematic diagramof the “classical” receiver for BD 23-25-27 GB can be seen in FIG. 1(but it should be noted that it has already advanced signal processingmeasures, like a Viterbi detector (VD)), and it will be appreciated by aperson skilled in the art that the “slicer” operates on raw samples ofthe HF signal waveform that is unequalised and asynchronous (thus, itneed not necessarily be synchronous).

Consider first traditional DC-control (based on the knownparity-preserve principle, or on the principle of merging-bits, or onthe principle of combi-codes) which aims to minimize the varianceσ_(RDS) ² of the running digital sum, denoted RDS. Which is defined as(with b_(i) the bipolar channel bits (also known as NRZI-bits) withvalues +1): $\begin{matrix}{{RDS}_{i} = {\sum\limits_{j = {- \infty}}^{i}b_{j}}} & (2)\end{matrix}$

The state-of-the-art DC-control is shown in FIG. 6. The DC controldecision points divide the bitstream in consecutive segments. For agiven segment, the channel bitstream is encoded by means of the RLLencoder in step (1) for both choices of DC-control (for instance in thecase of the parity-preserve method parity-bit equal to “1” or “0”, or,in the case of merging bits in the channel bitstream, merging bits (ford=1) equal to “00” or “10” (or “01”) for both polarity choices). Forboth choices, the RDS is computed in step (2), together with itsvariance over the segment under consideration. Finally, the DC-controlchoice with the lowest variance of the RDS (or, alternatively, theDC-control choice with the lowest RDS-value in absolute value, at theend of the segment is selected in step (3).

The state-of-the-art “slicer control”, that is the control of the slicerlevel (denoted SL) at the receiver side, is implemented according to(with HF_(k) being the k-th sample of the (asynchronously orsynchronously sampled) high-frequency signal waveform):SL_(k+1)=SL_(k)+εSign(HF_(k)−SL_(k))  (3)

At the signal transitions, the above equation can be refined so that itaccounts for phase-offsets relative to the centre point between twosampling points (for simplicity, we have omitted this refinement). Theparameter ε determines the update speed of the slicer-level control (asmaller value of ε corresponds with a slower update rate, and thus asmaller bandwidth in the adaptation). FIG. 7 shows the principle ofSL-control for a large value of ε (this value has been madeunrealistically large just for display purposes in the figure; practicalvalues of e lie in the range 10⁻³ to 10⁻⁵). The result of the SL-controlis that the slicer level equals the median of the HF signal waveform:the number of HF samples above the slicer level is equal to the numberof samples below the slicer level (within a window of a minimum numberof samples, that is determined via the update rate ε).

The underlying assumption of the SL-control in Eq. (3) is that simplethreshold detection yields a very good bER-performance (for asynchronous HF signal waveform, the Sign operation in Eq. (3) is nothingelse than threshold detection). Note that the decision of the thresholddetector yields only a real bit-decision in the case of the synchronoussignal waveform (since there is no one-to-one correspondence between theasynchronous samples and the channel bits).

The assumption that threshold detection yields a good bER is only validfor capacities below 25 GB (for a 12 cm disc using a BD optical pick-upunit, OPU); for increasingly higher capacities, this assumption becomesmore and more unreliable, and therefore its lack of validity will affectthe performance of the slicer-level control in a negative sense. Becauseof the erroneous decisions made by the threshold detection at highcapacities, the SL-control in the receiver does just the opposite actionof what it was supposed to do by the DC-control at the encoder side.

The key of the problem is the mismatch of the exact bits used forDC-control in Eq. (2) and the decision errors by the slicing operation(Sign) as used in the SL-control of Eq. (3).

A logical solution would be to use better bit-decisions for theSL-control: those better bit-decisions must then come from aViterbi-like bit-detector which performs sequence detection using themulti-valued character of the signal waveform for a controlled ISI.However, a first disadvantage of this solution is that thosebit-decisions are produced by the Viterbi detector with a non-negligibledelay (e.g. due to the trace-back operation of the Viterbi) whichamounts to, for example, 5 times the extent in bits of the channel'simpulse response function. A second disadvantage is that the HF samplesat the inut of the Viterbi detector suffer from DC-offsets (as they aredue to fingerprints etc) which may lead to a worse performance of thebit-detector.

The solution offered by the present invention eliminates the DC-offsetsby an adapted version of the traditional slicer. Thus it is necessary toeliminate the mismatch between the decisions in the slicer adaption andthe exact bits as they are used in the DC-control. This mismatch becomesmore significant with increasing capacity.

The solution offered by this exemplary embodiment of the presentinvention is to perform DC-control at the encoder, hereby controllingthe running-digital-sum, not based on the exact bits, but based onthreshold decisions from a synthetic HF signal waveform (denotedHF_(syn)) that is obtained from a nominal MTF (modulation transferfunction) or its IRF (impulse response function; denoted h_(k)). Thenominal IRF must be matched to the capacity for which the storageapplication is meant for (e.g. by the cut-off frequency Ω_(c) of theBraat-Hopkins MTF in optical recording).

For the simple case of a linear channel, the synthetic HF signalwaveform is simply obtained by the convolution sum (with b_(k) thebipolar channel bits):HF_(syn,k)=Σ_(m)h_(m)b_(k-m.)  (4)

The case of channel non-linearities (e.g. with pit-land asymmetry) willbe dealt with in a later section. Based on this synthetic waveform, weintroduce a modified RDS parameter (denoted $\begin{matrix}{{\overset{\sim}{\left. {RDS} \right)}\quad{by}\text{:}\quad{\overset{\sim}{RDS}}_{I}} = {\sum\limits_{j = {- \infty}}^{i}{{Sign}\left( {HF}_{{syn},j} \right)}}} & (5)\end{matrix}$

DC-control can then be performed by minimization of the variance σ_(RDS)²⁻. By doing so, the DC-control at the encoder anticipates on (most of)the bit-errors that will be made by a simple threshold detector in theSL-control module in the receiver, thus removing the mismatch referredto above.

The DC-control according to this exemplary embodiment is shown in FIG.8. There are two additional steps: step (4) comprises the computation ofthe synthetic HF signal waveform, and step (5) in which the thresholddetection on the synthetic waveform is applied, which yields theintermediate channel bits, which replace the exact bits in thestate-of-the-art approach of FIG. 6.

Note that the encoder requires knowledge of the nominal MTF of thechannel (which depends on the targeted density on the disc).

Thus HF signal waveforms have been simulated for the Braat-Hopkinsmodel. In the above example d=1, k=7 Jacobi code has been used with twomerging bits for DC-control (because of the simplicity of this RLLencoder). A DC-control segment of the channel bitstream comprises 66channel bits. NO DC-level variations are introduced, so we know that theDC-level should be exactly at zero. Then, the standard deviation of theslicer-level after slicer-level adaptation was computed with differentvalues of the update-rate parameter ε. The results for ε=0.001, ε=0.0001and ε=0.00001 are shown in FIG. 9, FIG. 10 and FIG. 11 respectively. Theimprovement of the new DC-control procedure, based on the modified RDScan be clearly seen: for ε=0.0001 at BD 35 GB, the improvement is equalto a factor of three.

In the case of non-negligible pit-land asymmetry, it is not possible torely on the HF-model of Eq. (4). Instead, the known A-parameter modelcan be used (see, for example, H. Poziolis, J. W. M. Bergmans, W. M. J.Coene, “Modeling and Compensation of Asymmetry in Optical Recording”,IEEE Transactions on Communications, Vol. 50, No. 12, December 2002, pp.2052-2063), in which an intermediate ternary bitstream, denoted {tildeover (b)}_(k) is defined as: $\begin{matrix}{{\overset{\sim}{b}}_{k} = {b_{k} - {\frac{1}{4}\left( {{A} + {Ab}_{k}} \right) \times {\left( {{2b_{k}} - b_{k + 1} - b_{k - 1}} \right).}}}} & (6)\end{matrix}$

The adapted HF signal waveform, denoted HF^(A), is then obtained as:HF _(syn,k) ^(A) =c+Σ _(m) {tilde over (b)} _(k-m.)  (7)where c is a corrective DC-term such that the middle of the inner eyecorresponds with level ‘0’ (as is required if we want to “slice” atlevel ‘0’ for the bit-decisions used for DC-control); it is given by:$\begin{matrix}{c = {\frac{A}{2}{\sum\limits_{m}{h_{m}.}}}} & (8)\end{matrix}$

Note that the encoder must now not only know the nominal MTF, but alsothe pit-land asymmetry parameter A. Moreover, the encoder has to set anabsolute polarity with which the channel bitstream will be written todisc, because the asymmetry does not allow an overall polarity inversionof the channel bitstream.

The actual slicer level that results from the slicer-control loop isusually the centre of the inner eye. However, it is also possible to setthe slicer level to a different signal level, by applying a certainDC-offset before the Sign-operation in Eq. (5).

Thus, in summary and as explained above, the loss of performance of theslicer adaptation at high capacities is due to the mismatch between theexact bits used in the computation of the RDS for the DC-control on theone hand (as illustrated in FIG. 12), and the often erroneous thresholddecisions based on the HF waveform on the other hand. This mismatch canbe resolved by the present invention by performing a new method ofDC-control at the encoder: the RDS is modified such that it is not basedon the exact channel bits, but on intermediate channel bits that areobtained as the threshold decisions from a synthetic HF signal waveformthat is generated based on a nominal MTF (modulation transfer function)or its IRF (impulse response function) of the channel. In this way, theimpact of the erroneous threshold decisions in the receiver are alreadytaken into account at the encoder, and the slicer control is no longernegatively affected thereby. Note that, in the proposed method, theencoder needs to be informed of the nominal MTF or IRF of the channel.Extensions to channels with non-negligible pit-land symmetry are alsoproposed.

It should be noted that the above-mentioned embodiment illustratesrather than limits the invention, and that those skilled in the art willbe capable of designing many alternative embodiments without departingfrom the scope of the invention as defined by the appended claims. Inthe claims, any reference signs placed in parentheses shall not beconstrued as limiting the claims. The word “comprising” and “comprises”,and the like, does not exclude the presence of elements or steps otherthan those listed in any claim or the specification as a whole. Thesingular reference of an element does not exclude the plural referenceof such elements and vice-versa. The invention may be implemented bymeans of hardware comprising several distinct elements, and by means ofa suitably programmed computer. In a device claim enumerating severalmeans, several of these means may be embodied by one and the same itemof hardware. The mere fact that certain measures are recited in mutuallydifferent dependent claims does not indicate that a combination of thesemeasures cannot be used to advantage.

1. Apparatus for creating a substantially DC-controllable channelbitstream based upon a channel code, representative of a modulatedsignal received via a channel, the apparatus comprising means forreceiving data representative of a nominal modulation transfer functionor impulse response function of said channel, means for generating asynthetic high frequency signal waveform using said nominal modulationtransfer function or impulse response function, means for performingthreshold detection in respect of said synthetic high frequency signalwaveform to produce intermediate channel bits, and means for computing arunning digital sum using said intermediate channel bits.
 2. Apparatusaccording to claim 1, wherein the code representative of the modulatedsignal is a run length limited (RLL) code.
 3. Apparatus according toclaim 2, wherein the apparatus includes means for performing RLLencoding in respect of a plurality of user bits that are to be encoded.4. Apparatus according to claim 3, wherein said RLL encoding ie peformedprior to computation of the running digital sum.
 5. An encoder includingapparatus according to claim
 1. 6. An optical data storage system,comprising an encoder according to claim 5, and a receiver comprisingslicer apparatus for performing threshold detection in respect of a highfrequency signal waveform derived from a modulated signal to create adigital signal representative thereof.
 7. A system according to claim 6,the receiver further comprising slicer-control means for updating athreshold value in respect of the slicer apparatus in response tochanges in said high frequency signal waveform, so as to correct for aDC-offset in said high-frequency signal waveform.
 8. A system accordingto claim 6, the receiver further including a sequence detector for saidhigh frequency signal waveform.
 9. A system according to claim 8,wherein said sequence detector is a Viterbi sequence detector.
 10. Amethod for creating a substantially DC-controllable channel bitstreambased upon a channel code, representative of a modulated signal receivedvia a channel, the method comprising receiving data representative of anominal modulation transfer function or impulse response function ofsaid channel, generating a synthetic high frequency signal waveformusing said nominal modulation transfer function or impulse responsefunction, performing threshold detection in respect of said synthetichigh frequency signal waveform to produce intermediate channel bits, andcomputing a running digital sum using said intermediate channel bits.11. A receiver with a slicer apparatus for controlling the DC-level of areceived high frequency signal waveform, wherein said slicer apparatusperforms threshold decisions in respect of said received high frequencysignal waveform, and performs adjustements of the slicer-level inaccordance with said threshold decisions, wherein said signal waveformresults from a channel bitstream transmitted over the channel, that hasbeen encoded by means of the method of claim 10.